Quadrature mixer circuit including three-input local mixers

ABSTRACT

In a quadrature mixer circuit for receiving a radio frequency signal to generate first and second quadrature output signals, a first three-input mixer receives the radio frequency signal, a first local signal having a first frequency and a second local signal having a second frequency to generate the first quadrature output signal, and a second three-input mixer receives the radio frequency signal, the first local signal and the second local signal to generate the second quadrature output signal. The second local signal received by the first three-input mixer and the second local signal received by the second three-input mixer being out of phase by π/2 from each other.

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] The present invention relates to a quadrature mixer circuitapplied to a wireless receiver, and more particularly, to a quadraturemixer circuit applied to a zero intermediate frequency (IF) type (aso-called direct conversion type) wireless receiver or a low IF typewireless receiver.

[0003] 2. Description of the Related Art

[0004] While super heterodyne type wireless receivers have excellentnoise figure (NF) characteristics, the super heterodyne type wirelessreceivers have a large number of components including local oscillators,image removing filters and an IF band-pass filter, which is an obstaclefor incorporating a radio frequency (RF) portion and a baseband portioninto one chip.

[0005] In order to decrease the number of components, various directconversion type wireless receivers have been developed. In this case,the improvement of quadrature mixer circuits applied to such directconversion type wireless receivers is indispensable.

[0006] In a first prior art quadrature mixer circuit using two-inputmixers (see: FIG. 29 of JP-A-9-205382), since a local oscillator signalhas the same frequency as that of a radio frequency (RF) signal, a DCoffset cannot be completely removed, which requires DC offset removingcircuits. Also, trouble in reception sensitivity may be generated.Further, the reception sensitivity of other wireless receivers may besuppressed. This will be explained later in detail.

[0007] Even in a second prior art quadrature mixer circuit usingtwo-input mixers and a local oscillator signal having a half frequencyof the RF signal, the same disadvantages as the first prior artquadrature mixer circuit exist. This also will be explained later indetail.

[0008] In a third prior art quadrature mixer circuit (see: JP-A-9-205382& Takafumi Yamaii et al, “An I/Q Active Balanced Harmonic Mixer with IM2Cancelers and a 45° Phase Shifter”, IEEE Journal of Solid-StateCircuits, Vol. 33, No. 12, pp. 2240-2246, December 1998), two-inputeven-ordered harmonic mixers, a voltage controlled oscillator having afrequency different from the frequency of the RF signal and a π/4 phaseshifter are provided. In the third prior art quadrature mixer circuit,however, it is difficult to realize the π/4 phase shifter. This alsowill be explained later in detail.

SUMMARY OF THE INVENTION

[0009] It is an object of the present invention to provide a quadraturemixer circuit capable of decreasing the DC offset, suppressing thereception trouble without a π/4 phase shifter and decreasing the numberof components such as IF filters and second filters.

[0010] According to the present invention, in a quadrature mixer circuitfor receiving an RF signal to generate first and second quadratureoutput signals, a first three-input mixer receives the RF signal, afirst local oscillator signal having a first frequency and a secondlocal oscillator signal having a second frequency to generate the firstquadrature output signal, and a second three-input mixer receives the RFsignal, the first local oscillator signal and the second localoscillator signal to generate the second quadrature output signal. Thesecond local oscillator signal received by the first three-input mixerand the second local oscillator signal received by the secondthree-input mixer are out of phase by π/2 from each other.

BRIEF DESCRIPTION OF THE DRAWINGS

[0011] The present invention will be more clearly understood from thedescription set forth below, as compared with the prior art, withreference to the accompanying drawings, wherein;

[0012]FIG. 1 is a block circuit diagram illustrating a first prior artquadrature mixer circuit applied to a direct conversion type wirelessreceiver;

[0013]FIG. 2 is a circuit diagram for explaining a down-conversionsystem of the direct conversion type wireless receiver of FIG. 1;

[0014]FIG. 3 is a circuit diagram for explaining a DC offset by theself-mixing phenomenon in FIG. 1;

[0015]FIG. 4 is a circuit diagram illustrating a second prior artquadrature mixer circuit applied to a direct conversion type wirelessreceiver;

[0016]FIG. 5 is a circuit diagram illustrating a third prior artquadrature mixer circuit applied to a direct conversion type wirelessreceiver;

[0017]FIG. 6 is a block circuit diagram illustrating a first embodimentof the quadrature mixer circuit applied to a direct conversion typewireless receiver according to the present invention;

[0018]FIG. 7 is a block circuit diagram illustrating a second embodimentof the quadrature mixer circuit applied to a direct conversion typewireless receiver according to the present invention;

[0019]FIG. 8A is a detailed circuit diagram of the post stage of theJohnson counter of FIG. 7;

[0020]FIG. 8B is a waveform diagram showing the input and output signalsof the post stage of the Johnson counter of FIG. 8A;

[0021]FIG. 9 is a circuit diagram of a doubly-polarity switching mixerused as each of the three-input mixers of FIGS. 6 and 7;

[0022]FIG. 10A is a detailed circuit diagram of the doubly-polarityswitching mixer of FIG. 9;

[0023]FIG. 10B is a detailed circuit diagram of the linear differentialcircuit of FIG. 10A;

[0024]FIG. 11A is another detailed circuit diagram of thedoubly-polarity switching mixer of FIG. 9;

[0025]FIG. 11B is a detailed circuit diagram of the linear differentialcircuit of FIG. 11A;

[0026]FIG. 12 is a circuit diagram illustrating a modification of thedoubly-polarity switching mixer of FIG. 10A;

[0027]FIG. 13 is a circuit diagram illustrating a modification of thedoubly-polarity switching mixer of FIG. 11A;

[0028]FIG. 14A is a circuit diagram illustrating a modification of thedoubly-polarity switching mixer of FIG. 12;

[0029]FIG. 14B is a detailed circuit diagram of the linear differentialcircuit of FIG. 14A;

[0030]FIG. 15A is a circuit diagram illustrating a modification of thedoubly-polarity switching mixer of FIG. 13;

[0031]FIG. 15B is a detailed circuit diagram of the linear differentialcircuit of FIG. 15A;

[0032]FIG. 16 is a block circuit diagram illustrating a third embodimentof the quadrature mixer circuit applied to a direct conversion typewireless receiver according to the present invention;

[0033]FIG. 17 is a block circuit diagram illustrating a fourthembodiment of the quadrature mixer circuit applied to a directconversion type wireless receiver according to the present invention;and

[0034]FIGS. 18 and 19 are tables showing the frequency division numbersof the frequency dividers of FIG. 17.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0035] Before the description of the preferred embodiments, prior artquadrature phase shift circuits applied to a direct conversion typewireless receiver will be explained with reference to FIGS. 1, 2, 3, 4and 5.

[0036] In FIG. 1, which illustrates a first prior art quadrature phaseshift circuit applied to a direct conversion type wireless receiver(see: FIG. 29 of JP-A-9-205382), an antenna 1 is connected via a lownoise amplifier 2 to a quadrature mixer circuit 3 which is formed bytwo-input mixers 31 and 32, a voltage controlled oscillator 33 and a π/2phase shifter 34.

[0037] Also, the low frequency components of the output signals of thetwo-input mixers 31 and 32 pass through low-pass filters 4 and 5,respectively, and then, the gains of the output signals of the low-passfilters 4 and 5 are controlled by automatic gain control (AGC)amplifiers 6 and 7, respectively. Thus, baseband components I and Q areobtained.

[0038] Further, the baseband components I and Q are subjected toanalog-to-digital conversion by analog/digital (A/D) converters 8 and 9,and then are supplied to a digital signal processor (DSP) 10 serving asa demodulator.

[0039] The principle of the two-input mixers 31 and 32 is explainedbelow.

[0040] In the two-input mixers 31 and 32, a second-order term of thetransfer characteristics of a non-linear element is used. In thenon-linear element, an input u and an output f(u) are represented by

f(u)=a ₀ +a ₁ u+a ₂ u ² + . . . +a _(n) u ^(n)+  (1)

[0041] When an RF signal uRV having a frequency f_(RF) and a localoscillator signal u_(Lo). having a frequency f_(Lo) are mixed at thenon-linear element, the second term a₂u^(z) of the formula (1) isrepresented by

a _(z) u ² =a ₂(u_(RF) +u _(LO))² =a ₂(u _(RF) ² +u _(LO) ²+2u _(RF) u_(LO))  (2)

[0042] In this case,

u _(RF) =u _(RF)·cos(2πf _(RF) t)  (3)

u _(LO) =U _(LO)·cos(2πf _(LO) t)  (4)

[0043] Therefore, the third term of the formula (2) is represented by

2U _(RF) ·u _(LO)=2U _(RF) ·U _(LO)·cos(2πf _(RF) t)·cos(2πf _(LO) t)=U_(RF) ·U _(LO)·[cos{2π(f _(RF) −f _(LO))t}+cos{2π(f _(RF) +f_(LO))t}]  (5)

[0044] In such a down-conversion system as illustrated in FIG. 1 wherethe two-input mixer 32 is followed by the low-pass filter 5, as shown inFIG. 2, only the low frequency componentU_(RF)·U_(LO)·cos{2π(f_(RF)−f_(LO))t} in the formula (5) is obtained.Note that a difference frequency |f_(RF)−f_(LO)| is defined as anintermediate frequency f_(IF).

[0045] If an ideal mixer consisting of a multiplier has the sameconversion gain in both frequencies far and f_(RF)−f_(LO)) the in-bandnoise power converted from two bands around the both frequencies f_(RF)and f_(RF)−f_(LO) is twice (=1/(1²+1²)) from the formula (5). Therefore,the noise figure (NF) of the ideal mixer becomes damaged by two, i.e.,3.01 dB, as compared with a circuit with a single-input signal and asingle-output signal.

[0046] On the other hand, since the local oscillator signal ULO issupplied via the π/2 phase shifter 34 to the two-input mixer 31, a localoscillator signal U_(LO)′ supplied to the two-input mixer 31 isrepresented by

u _(LO) ′=U _(LO)·sin(2πf _(LO) t)  (6)

[0047] Therefore, the third term of the formula (2) is represented by

2U _(RF) ·u _(LO)′=2U _(RF) ·U _(LO)·cos(2πf _(RF) t)·sin(2πf _(LO) t)=U_(RF) ·U _(LO)·[−sin{2π(f _(RF) −f _(LO))t}+sin{2π(f _(RF) +f_(LO))t}]  (7)

[0048] Thus, in a down-conversion system as illustrated in FIG. 1 wherethe two-input mixer 31 is followed by the low-pass filter 4, thecomponent, U_(RF)·U_(LO)·sin{2π(f_(RF)−f_(LO))t} is obtained. Note thata difference |f_(RF)−f_(LO)| is also defined as an intermediatefrequency f_(IF).

[0049] In the direct conversion type wireless receiver of FIG. 1,f_(RF)=f_(LO), so that there is no intermediate frequency f_(IF)(=0).

[0050] The direct conversion type wireless receiver of FIG. 1 has thefollowing advantages.

[0051] {circle over (1)} Since there is no intermediate frequencyf_(IF)(=0), no suppression of image is necessary, that is, no imageremoving filters are necessary. Also, there are few sources forgenerating spurious waves. Note that the low-pass filters 4 and 5 areprovided in a baseband portion, and therefore, the low-pass filters 4and 5 are easily incorporated into an integrated circuit.

[0052] {circle over (2)} At The baseband portion as well as an RFportion including the low noise amplifier 2 and the quadrature mixercircuit 3 can be easily incorporated into one chip.

[0053] Thus, the direct conversion type wireless receiver of FIG. 1 canbe easily incorporated into one chip, which would decrease the size anddecrease the manufacturing cost.

[0054] On the other hand, the direct conversion type wireless receiverof FIG. 1 has the following disadvantages.

[0055] {circle over (1)} A DC offset may be generated. That is, a slightdifference between the frequency f_(RF) of the RF signal u_(RF) and thefrequency f_(LO) of the local oscillator signal u_(LO) appear as a DCoffset at the output signal of each of the two-input mixers 31 and 32.Also, as illustrated in FIG. 3, the local oscillator signal u_(LO) leaksfrom one input of the two-input mixer 32 (31) to the other input of thetwo-input mixer 32 (31). As a result, the leaked local oscillator signalis reflected at the output of the low noise amplifier 2 and at theantenna 1 as indicated by X1 and X2, respectively. The reflected localoscillator signals X1 and X2 are down-converted and result in a DCoffset which is called a DC offset caused by the self-mixing phenomenon.Since the above-mentioned DC offset is not stable, the DC offset servesas a low frequency noise. In order to remove the DC offset, DC offsetremoving circuits are required, which is an obstacle in realizing onechip wireless receiver.

[0056] {circle over (2)} Ad Since the frequency f_(LO) of the localoscillator signal u_(LO) coincides with the frequency f_(RF) of the RFsignal, the leakage of the local oscillator signal u_(LO) within thewireless receiver of FIG. 1 generates trouble in reception sensitivityaround the frequency f_(RF).

[0057] {circle over (3)} Antenna radiation of the local oscillatorsignal u_(LO) suppresses the reception sensitivity of other wirelessreceivers receiving RF signals using approximately the same frequencyf_(LO).

[0058] In FIG. 4, which illustrates a second prior art quadrature mixercircuit applied to a direct conversion type wireless receiver, a¼-frequency divider 35, a two-input mixer 36 and a band-pass filter 37are added to the elements of FIG. 1. That is, if the local oscillatorsignal u_(LO) of the voltage controlled oscillator 33 is f_(LO), a localoscillator signal having a frequency (¾) f_(LO) is generated by the¼-frequency divider 35, the two-input mixer 36 and the band-pass filter37. Therefore,

f _(RF)=(¾)f _(LO)

[0059] As a result, the frequency f_(LO) of the voltage controlledoscillator 33 is different from the frequency f_(RF) of the RF signal.

[0060] Even in the direct conversion type wireless receiver of FIG. 4,since the frequency (¾) f_(LO) at an input of the two-input mixer 31(32)is the same as that at the frequency f_(RF) at another input of thetwo-input mixer 31(32), there are the same disadvantages as in thedirect conversion type wireless receiver of FIG. 1.

[0061] In FIG. 5, which illustrates a third prior art quadrature mixercircuit applied to a direct conversion type wireless receiver (see:JP-A-9-205382) & Takafumi Yamaji et al, “An I/Q Active Balanced HarmonicMixer with IM2 Cancelers and a 45° Phase Shifter”, IEEE Journal ofSolid-State Circuits, Vol. 33, No. 12, pp. 2240-2246, December 1998),the quadrature mixer circuit 3 is constructed by two-input even-harmonicmixers 31′ and 32′, a voltage controlled oscillator 33′ and a π/4 phaseshifter 34′.

[0062] The principle of the two-input even-harmonic mixers 31′ and 32′is explained below.

[0063] In the two-input even-harmonic mixers 31′ and 32′, a third-orderterm of the transfer characteristics of a non-linear element is used.

[0064] When an RF signal u_(RF) having a frequency f_(RF) and a localoscillator signal u_(LO) having a frequency f_(LO) are mixed at thenon-linear element, the third-order term a₃u³ of the formula (1) isrepresented by

a ₃ u ³ =a ₃(u _(RF) +u _(LO))³ =a ₃(u _(RF) ³ +u _(LO) ³+3u _(RF) ² u_(LO)+3u _(RF) u _(LO) ²)  (8)

[0065] The fourth term of the formula (8) is represented by

[0066] 3u _(RF) U _(LO) ²=3U _(RF) ·U _(LO) ²·cos(2πf _(RF) t)cos²(2πf_(LO) t)=3_(U) _(RF) ·U _(LO) ²cos(2πf _(RF) t){1+cos(4πf _(LO) t)}/2=3U_(RF) ·U _(LO) ²·{cos(2πf _(RF) t)+cos(2πf _(RF) t)cos(4πf _(LO)t)}/2=3U _(RF) ·U _(LO) ²[2 cos(2πf _(RF) t)+cos{2π(f _(RF) t−2f_(LO))t}+cos{2π(f _(RF) t+2f _(LO))t}]/4   (9)

[0067] In such a down-conversion system of FIG. 5 where the two-inputeven-harmonic mixer 32′ is followed by the low-pass filter 5, only thelow frequency component 3U_(RF)·U_(LO) ² cos{2π(f_(RF)t−2f_(LO))t} inthe formula (9) is obtained. Note that a difference |f_(RF)−2f_(LO)| isdefined as an intermediate frequency f_(IF).

[0068] If an ideal even-harmonic mixer has the same conversion gain inboth frequencies f_(RF), f_(RF)−2f_(LO) and f_(RF)+2f_(LO), the in-bandnoise power converted from the three bands around the frequenciesf_(RF), f_(RF)−2f_(LO) and f_(RF)+2f_(LO) is six-times (=1/(2²+1²+1²))from the formula (9) by squaring each amplitude thereof. Therefore, thenoise figure (NF) of the ideal even-harmonic mixer becomes damaged bysix, i.e., 7.78 dB, as compared with a circuit with a single-inputsignal and a single-output signal.

[0069] On the other hand, since the local oscillator signal ULO issupplied via the π/4 phase shifter 34′ to the two-input even-harmonicmixer 31′, a local oscillator signal U_(LO)′ supplied to the two-inputeven-harmonic mixer 31′ is represented by

u _(LO) ′u _(LO)·cos(2πf _(LO) t+π/4)  (10)

[0070] Therefore, the fourth-order term of the formula (8) isrepresented by

[0071] 3u _(RF) u _(LO)′²=3U _(RF) ·U _(LO) ² cos(2πf _(RF) t)cos²(2πf_(LO) t+π/4)=3U _(RF) ·U _(LO) ² cos(2πf _(RF) t){1+cos(4πf _(LO)t+π/2)}/2=3U _(RF) ·U _(LO) ²·cos(2πf _(RF) t){1−sin(4πf _(LO) t)}/2=3U_(RF) ·U _(LO) ²{cos(2πf _(RF) t)−cos(2πf _(RF) t)sin(4=πf _(LO)t)}/2=3U _(RF) ·U _(LO) ²·[2 cos(2πf _(RF) t)+sin{2π(f _(RF) t−2f_(LO))t}+sin{2π(f _(RF) t+2f _(LO))t}]/4  (11)

[0072] Thus, in a down-conversion system of FIG. 5 where the two-inputeven-harmonic mixer 31′ is followed by the low-pass filter 4, thecomponent 3U_(RF)·U_(LO) ²·sin{2π(f_(RF)t−2f_(LO))t} of the formula (14)is obtained. Note that a difference |f_(RF)t−2f_(LO)| is also defined asan intermediate frequency f_(IF).

[0073] In the quadrature mixer circuit 3 of FIG. 5, however, as statedin JP-A-9-205382, it is difficult to realize the π/4 phase shifter 34′.

[0074] In FIG. 6, which illustrates a first embodiment of the quadraturemixer circuit applied to a direct conversion type wireless receiveraccording to the present invention, the two-input mixers 31 and 32 ofFIG. 1 are replaced by three-input mixers 31″ and 32″, respectively, andthe voltage controlled oscillator 33 of FIG. 1 is replaced by twovoltage controlled oscillators 33″A and 33″B, which generate localoscillator signals u_(LO1), and u_(LO2) having frequencies f_(LO1) andf_(LO2), respectively. Note that each of the three-input mixers 31″ and32″ can be constructed by a three-input multiplier (see:JP-A-10-105632).

[0075] In the three-input mixers 31″ and 32″, a third-order term of thetransfer characteristics of a non-linear element is also used.

[0076] When the RF signal U_(RF) having the frequency f_(RF) and thelocal oscillator signals U_(LO1) and u_(LO2) having frequencies f_(LO1)and f_(LO2) are mixed at the non-linear element, the third-order terma₃u³ of the formula (1) is replaced by

[0077]a ₃ u ³ =a ₃(u _(RF) +u _(LO1) +u _(LO2))³ =a ₃(u _(RF) ³ +u_(LO1) ³ +u _(LO2) ³+3u _(RF) ² u _(LO1)+3u _(RF) ² u _(LO2)+3u _(RF) u_(LO1) ²+3u _(LO1) ² u _(LO2) +3u _(RF) u _(LO2) ²+3u _(LO1) u _(LO2)²+6u _(RF) u _(LO1) u _(LO2))  (13)

[0078] In this case, the RF signal u_(RF) and the local oscillatorsignals u_(LO1) and u_(LO2) are represented by

u _(RF) =u _(RF)·cos(2πf _(RF) t)  (14)

u _(LO1) =u _(LO1)·cos(2πf _(LO1) t)  (15)

u _(LO2) =u _(LO2)·cos(2πf _(LO2) t)  (16)

[0079] Generally, the following triple product of trigonometricfunctions is known;

cos α cos β cos γ={cos(α+β−γ)+cos(β+γ−α)+cos(γ+α−β)+cos(α+β+γ)}/4  (17)

[0080] Therefore, the tenth term of the formula (13) is represented by

[0081] 6u _(RF) u _(LO1) u _(LO2)=6U _(RF) ·U _(LO1) ·U _(LO2)·cos(2πf_(RF) t)cos(2πf _(LO1) t)cos(2πf _(LO2) t)=3U_(RF) ·U _(LO1) ·U_(LO2)·[cos{2π(f _(RF) +f _(LO1) −f _(LO2))t}+cos{2π(−f _(RF) +f _(LO1)+f _(LO2))t}+cos{2π(f _(RF) −f _(LO1) +f _(LO2))t}+cos{2π(f _(RF) +f_(LO1) +f _(LO2))t}]/2  (18)

[0082] In such a down-conversion system of FIG. 6 where the three-inputmixer 32″ is followed by the low-pass filter 5, only the low frequencycomponent 3U_(RF)·U_(LO1)·U_(LO2) cos{2π(−f_(RF)+f_(LO1)+f_(LO2))t} inthe equation (18) is obtained. Note that a difference|f_(RF)−f_(LO1)−f_(LO2)| is defined as an intermediate frequency f_(RF).

[0083] On the other hand, since the local oscillator signal u_(LO2) issupplied via the π/2 phase shifter 34 to the three-input mixer 31″, alocal oscillator signal u_(LO2)′ supplied to the three-input mixer 31″is represented by

u _(LO2) ′=U _(LO2) sin(2πf _(LO2) t)  (19)

[0084] Generally, the following triple product of trigonometricfunctions is known:

sin α cos β cos γ={sin(α+β−γ)+sin(β+γ−α)+sin(γ+α−β)−sin(α+β+γ)}/4  (20)

[0085] Therefore, the tenth term of the formula (13) is represented by

[0086] 6u _(RF) u _(LO1) u _(LO2)=6U _(F) ·U _(LO1) ·ULO2 ·cos(2πf _(RF)t)cos(2πf _(LO1) t)sin(2πf _(LO2) t)=3U _(RF) ·U _(LO1) ·U_(LO2)·[sin{2π(f _(RF) +f _(LO1) −f _(LO2))t}+sin{2π(−f _(RF) +f _(LO1)+f _(LO2))t}+sin{2π(f _(RF) −f _(LO1) +f _(LO2))t}+sin{2π(f _(RF) +f_(LO1) +f _(LO2))t}]/2  (21)

[0087] Thus, in a down-conversion system of FIG. 6 where the three-inputmixer 31″ is followed by the low-pass filter 4, the component3U_(RF)·U_(LO1)·U_(LO2)·sin{2π(f_(RF)+f_(LO1)+f_(LO2))t} of the formula(21) is obtained. Note that a difference |f_(RF)−f_(LO1)−f_(LO2)| isalso defined as an intermediate frequency f_(IF).

[0088] In FIG. 7, which illustrates a second embodiment of thequadrature mixer circuit applied to a direct conversion type wirelessreceiver according to the present invention, the voltage controlledoscillators 33″A and 33″B and the π/2 phase shifter 34 of FIG. 6 arereplaced by a voltage controlled oscillator 33″ and a Johnson counterformed by two ½-frequency dividers 71 and 72. In this case, the Johnsoncounter generates local oscillator signals u_(LO/4) and u_(LO/4)′ out ofphase by π/2.

[0089] In the three-input mixers 31″ and 32″,sin{π2(f_(RF)−f_(LO1)−f_(LO2))t} of the formula (21) is also used.

[0090] When the RF signal U_(RF) having the frequency f_(RF) and thelocal oscillator signals u_(LO) and u_(LO/4) having frequencies f_(LO)and f_(LO/4) are mixed at the non-linear element, the third-order terma₃u³ of the formula (1) is replaced by

[0091]a ₃ u ³ =a ₃(u _(RF) +u _(LO) +u _(LO/4))³ =a ₃(u _(RF) ³ +u _(LO)³ +u _(LO/4) ³+3u _(RF) ² u _(LO)+3u _(RF) ² u _(LO/4)+3u _(RF) u _(LO)²+3u _(LO2) u _(LO/4)+3u _(RF) u _(LO/4) ²3u _(LO) u _(LO/4) ²+6u _(RF)u _(LO) u _(LO/4))  (22)

[0092] In this case, the local oscillator signal u_(LO/4) is representedby

u _(LO/4) =u _(LO/4)cos{2π(f _(LO/4))t}  (23)

[0093] Therefore, the tenth term of the formula (22) is represented by

[0094] 6u _(RF) u _(LO) u _(LO/4)=6U _(RF) ·U _(LO) ·U _(LO/4)·cos(2πf_(RF) t)cos(2πf _(LO) t)cos{2π(f _(LO)/4)t}3U _(RF) ·U _(LO) ·U_(LO/4)·[cos{2π(f ^(RF)−5f _(LO)/4)t}+cos{2π(f _(RF)−3f_(LO)/4)t}+cos{2π(f _(RF)+5f _(LO)/4)t}+cos{2π(f _(RF)+3f_(LO)/4)t}]/2  (24)

[0095] In such a down-conversion system of FIG. 7 where the three-inputmixer 32″ is followed by the low-pass filter 5, only the low frequencycomponent 3U_(RF)·U_(LU)·U_(LO/4)·cos{2π(−f_(RF)+5f_(LO)/4)t} in theequation (24) is obtained. Note that a difference |f_(RF)−5f_(LO)/4| isdefined as an intermediate frequency f_(RF).

[0096] On the other hand, the local oscillator signal u_(LO/4)′ suppliedto the three-input mixer 31″ is represented by

u _(LO4) ′U _(LO/4)·sin{2π(f _(LO)/4)t}  (25)

[0097] Therefore, the tenth term of the formula (24) is represented by

[0098] 6u _(RF) u _(LO) u _(LO/4)=6U _(RF) ·U _(LO) ·U _(LO/4)·cos(2πf_(RF) t)cos(2πf _(LO) t)sin{2π(f _(LO)/4)t}=3U _(RF) ·U _(LO) ·U_(LO/4)[sin{2π(f _(RF)−5f _(LO)/4)t}+sin{2π(f _(RF)−3f_(LO)/4)t}+sin{2π(f _(RF)+5f _(LO)/4)t}+sin{2π(f _(RF)+3f_(LO)/4)t}]/2  (26)

[0099] Thus, in a down-conversion system of FIG. 7 where the three-inputmixer 31″ is followed by the low-pass filter 4, the component3U_(RF)·U_(LO1)·U_(LO2)·sin{2π(f_(RF)−5f_(LO)/4)t} of the formula (26)is obtained. Note that a difference |fRF−5f_(LO)/4| is also defined asan intermediate frequency f_(IF).

[0100]FIG. 8A is a detailed circuit diagram of the post stage of theJohnson counter of FIG. 7 and FIG. 8B is a waveform diagram showing theinput and output signals of the post stage of the Johnson counter ofFIG. 8A. That is, the post stage counter 72 is constructed by D-typeflip-flops connected in series, so that the local oscillator signalsU_(LO/4) and U_(LO/4)′ having the same frequencies f_(LO)/4(I) andf_(LO)/4(Q) are obtained from a signal having a frequency f_(LO)/2.

[0101] The three-input mixers 31″ and 32″ can be constructed bydoubly-polarity switching mixers instead of the three-input multipliers.Doubly-polarity switching mixers other than the three-input multipliersare easily integrated into one chip.

[0102] A typical doubly-polarity switching mixer will be explained nextwith reference to FIGS. 9, 10A, 10B, 11A, 11B, 12, 13, 14A, 14B, 15A and15B.

[0103] In FIG. 9, which is a circuit diagram of the doubly-polarityswitching mixer, an RF signal V_(RF)(t) is switched by two rectangularlocal oscillator signals S₁(t) and S₂(t). That is, switches 91, 92, 93and 94 controlled by the local oscillator signals S₁(t) and S₂(t)receive the RF signal V_(RF)(t) to generate an intermediate frequencysignal V_(IF). If the local oscillator signal S₁(t) has an amplitude of±U_(LO1) and a frequency of f_(LO1) and the local oscillator signalS₂(t) has an amplitude of ±U_(LO2) and a frequency of f_(LO2), S₁(t) andS₂(t) are represented by

S ₁(t)=U _(LO1)(4/π)[cos(2πf _(LO1) t)−(1/3)cos(6πf _(LO1)t)+(1/5)cos(10πf _(LO1) t)−(1/7)cos(14πf _(LO1) t)++ . . . ]  (27)

S ₂(t)=U _(LO2)(4/π)[cos(2πf _(LO2) t)−(1/3)cos(6πf _(LO2)t)+(1/5)cos(10πf _(LO2) t)−(1/7)cos(14πf _(LO2) t)+ . . . ]  (28)

[0104] Also, if S₁(t)=U_(LO1)sgn(S₁(t)) and S₂(t)=U_(LO2)sgn(S₂(t)),

V _(IF)(t)=u_(RF) −U _(LO1) sgn(S ₁(t)U _(LO2) sgn(S ₂(t))  (29)

u _(RF) =U _(RF) cos(2πf _(RF) t)  (30)

[0105] Therefore,

V _(IF)(t)=u _(RF) ·U _(LO1) sgn(S ₁(t))U _(LO2) sgn(S ₂(t))=u _(RF)cos(2πf _(RF) t)U _(LO1) sgn(S ₁(t))U _(LO2) sgn(S ₂(t))  (31)

[0106] Since sgn(S₁(t)) and sgn(S₂(t)) are −1and +1 and +1 and +1,respectively, their absolute values are represented by

|sgn(S ₁(t))|=1  (32)

|sgn(S ₂(t))|=1  (33)

[0107] Therefore, sgn(S₁(t)) and sgn(S₂(t)) are represented by

sgn(S ₁(t))=(4/π)[cos(2πf _(LO1) t)−(1/3)cos(6πf _(LO1) t)+(5/1)cos(10πf_(LO1) t)−(1/7)cos(14πf _(LO1) t)+ . . . ]  (34)

sgn(S ₂(t))=(4/π)[cos(2πf _(LO2) t)−(1/3)cos(6πf _(LO2) t)+(5/1)cos(10πf_(LO1) t)−(1/7)cos(14πf _(LO2) t)+ . . . ]  (35)

[0108] Thus, the formula (31) is represented by $\begin{matrix}\begin{matrix}{{V_{IF}(t)} = \quad {{u_{RF} \cdot U_{L01}}{{{sgn}\left( {S_{1}(t)} \right)} \cdot U_{L02}}{{sgn}\left( {S_{2}(t)} \right)}}} \\{= \quad {{U_{RF} \cdot U_{L01} \cdot U_{L02} \cdot \cos}\quad {\left( {2\quad \pi \quad f_{RF}t} \right) \cdot {{sgn}\left( {S_{1}(t)} \right)} \cdot {{sgn}\left( {S_{2}(t)} \right)}}}} \\{= \quad {\left( {16/\pi^{2}} \right){U_{RF} \cdot U_{L01} \cdot U_{L02} \cdot {{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}\left\lbrack {{\cos \left( {2\quad \pi \quad f_{L01}t} \right)} -} \right.}}}} \\{\quad {{\left( {1/3} \right){\cos \left( {6\quad \pi \quad f_{L01}t} \right)}} + {\left( {1/5} \right){\cos \left( {10\quad \pi \quad f_{L01}t} \right)}} -}} \\{\left. \quad {{\left( {1/7} \right){\cos \left( {14\quad \pi \quad f_{L01}t} \right)}} + \quad \cdots} \right\rbrack \cdot \left\lbrack {{\cos\left( {2\quad \pi \quad f_{L02}t} \right)} -} \right.} \\{\quad {{\left( {1/3} \right){\cos \left( {6\quad \pi \quad f_{L02}t} \right)}} + {\left( {1/5} \right){\cos \left( {10\quad \pi \quad f_{L02}t} \right)}} -}} \\\left. \quad {{\left( {1/7} \right){\cos \left( {14\quad \pi \quad f_{L02}t} \right)}} + \quad \cdots} \right\rbrack \\{= \quad {\left( {16/\pi^{2}} \right){U_{RF} \cdot U_{L01} \cdot U_{L02} \cdot}}} \\{\quad \left\lbrack {{{\cos\left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {2\quad \pi \quad f_{L01}t} \right)}{\cos \left( {2\quad \pi \quad f_{L02}t} \right)}} +} \right.} \\{\quad {{\left( {1/9} \right){\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {6\quad \pi \quad f_{L01}t} \right)}{\cos \left( {6\quad \pi \quad f_{L02}t} \right)}} +}} \\{\quad {{\left( {1/25} \right){\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {10\pi \quad f_{L01}t} \right)}{\cos \left( {10\pi \quad f_{L02}t} \right)}} +}} \\{\quad {{\left( {1/49} \right){\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {14\quad \pi \quad f_{L01}t} \right)}{\cos \left( {14\quad \pi \quad f_{L02}t} \right)}} -}} \\{\quad {\left( {1/3} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {6\quad \pi \quad f_{L01}t} \right)}{\cos \left( {2\pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {2\quad \pi \quad f_{L01}t} \right)}{\cos \left( {6\quad \pi \quad f_{L02}t} \right)}} \right\} +} \\{\quad {\left( {1/5} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {10\quad \pi \quad f_{L01}t} \right)}{\cos \left( {2\quad \pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {2\quad \pi \quad f_{L01}t} \right)}{\cos \left( {10\quad \pi \quad f_{L02}t} \right)}} \right\} -} \\{\quad {\left( {1/7} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {14\quad \pi \quad f_{L01}t} \right)}{\cos \left( {2\quad \pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {2\quad \pi \quad f_{L01}t} \right)}{\cos \left( {14\pi \quad f_{L02}t} \right)}} \right\} -} \\{\quad {\left( {1/15} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {6\quad \pi \quad f_{L01}t} \right)}{\cos \left( {10\quad \pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {10\quad \pi \quad f_{L01}t} \right)}{\cos \left( {6\quad \pi \quad f_{L02}t} \right)}} \right\} -} \\{\quad {\left( {1/21} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {6\quad \pi \quad f_{L01}t} \right)}{\cos \left( {14\pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {14\pi \quad f_{L01}t} \right)}{\cos \left( {6\quad \pi \quad f_{L02}t} \right)}} \right\} -} \\{\quad {\left( {1/35} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {10\quad \pi \quad f_{L01}t} \right)}{\cos \left( {14\quad \pi \quad f_{L02}t} \right)}} +} \right.}} \\\left. {\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {14\quad \pi \quad f_{L01}t} \right)}{\cos \left( {10\quad \pi \quad f_{L02}t} \right)}} \right\} + \cdots} \right\rbrack\end{matrix} & (36)\end{matrix}$

[0109] Apparent from the formula (36), when the RF signal U_(RF) isswitched by the polarities of the local oscillator signals S₁(t) andS₂(t), basic waves and harmonic waves of the three signals are obtained.

[0110] For example, four frequency components such ascos{2π(f_(RF)−f_(LO1)−f_(LO2))t}, cos{2π(f_(RF)−f_(LO1)+f_(LO2))t},cos{2π(f_(RF)+f_(LO1)+f_(LO2))t} and cos{2π(f_(RF)+f_(LO1−f) _(LO2))t}are obtained from the triple product cos(2πf_(RF)t)cos(2πf_(LO1)t)cos(2πf_(LO2)t) of basic waves with reference to the formula of tripleproduct of trigonometric functions shown in the formula (17).

[0111] In this case, if the doubly-polarity switching mixer of FIG. 9constitutes a down-conversion system where a low-pass filter followsthis doubly-polarity switching mixer, only the low frequency componentcos{2π(f_(RF)−f_(LO1)−f_(LO2))t} is obtained. Note that a difference|f_(LO1)−f_(LO2)| is defined as an intermediate frequency f_(IF).

[0112] Also, four frequency components such ascos[2π{f_(RF)−(2m+1)f_(LO1)−(2m′+1)f_(LO2)}t],cos[2π{f_(RF)(2m+1)f_(LO1)+(2m′+1)f_(LO2)}t],cos[2π{f_(RF)+(2m+1)f_(LO1)+(2m′+1)f_(LO2)}t] andcos[2π{F_(RF)+(2m+1)f_(LO1)−(2m′+1)f_(LO2)}t] are obtained from tripleproducts cos(2πf_(RF)t) cos{2π(2m+1)f_(LO1)t}cos{2π(2m′+1)f_(LO2)t} (m,m′=1, 2, . . . ) of the RF signal V_(RF)(t) and the odd-higher harmonicwaves of the local oscillator signals S1(t) and S₂(t) with reference tothe formula of triple product of trigonometric functions shown in theformula (17). In this case, these triple products decay with acoefficient of 1/{(2m+1) (2m′+1)}. Also, the frequencies of these tripleproducts are on the odd-higher order. For example, whenf_(LO1)=2nf_(LO2) (n=2, 3, . . . ) and f_(RF)=f_(LO1)+f_(LO2), theclosest frequency is 7f_(LO1)/4. Here, if f_(LO2)=f_(LO1)/2 and m=m′=1,3f_(LO1)/2=f_(RF).

[0113] If an ideal doubly-polarity switching mixer has the sameconversion gain in frequencies f_(RF)±if_(LO1)±jf_(LO2), the in-bandnoise power converted from all bands around the frequenciesf_(RF)±if_(LO1)±jf_(LO2) is about six-times (=4(1/1²+1/3²+1/5²+ . . .)(1/2²+1/3²+1/5²+ . . . )=π⁴/16=6.088) from the formula (36). Note1/1²+1/3²+1/5²+ . . . +1/(2i+1)²+ . . . =π²/8. Therefore, the noisefigure (NF) of the ideal doubly-polarity switching mixer becomes damagedby 6.088, i.e., 7.844 dB, as compared with a circuit with a single-inputsignal and a single-output signal.

[0114] On the other hand, when a local oscillator signal S₂′(t) out ofphase by π/2 from the local oscillator signal S₂(t) instead of the localoscillator signal S₂(t) is input to the doubly-polarity switching mixerof FIG. 9, the formulae (27), (28), (34) and (35) are replaced by

S ₁(t)=U _(LO1)(4/π)[cos(2πf _(LO1) t)−(1/3)cos(6πf _(LO1)t)+(1/5)cos(10πf _(LO1) t)−(1/7)cos(14πf _(LO1) t)+ . . . ]  (37)

S ₂(t)′U _(LO2)(4/π)[sin(2πf _(LO2) t)+(1/3)sin(6πf _(LO2)t)+(1/5)sin(10πf _(LO2) t)+(1/7)sin(14πf _(LO2) t)+ . . . ]  (38)

sgn(S ₁(t))=(4/7)[cos(2πf _(LO1) t)−(1/3)cos(6πf _(LO1) t)+(5/1)cos(10πf_(LO1) t)−(1/7)cos(14πf _(LO1) t)+ . . . ]  (39)

sgn(S ₂(t)′)=(4/π)[−sin(2πf _(LO2) t)+(1/3)sin(6πf _(LO2)t)−(5/1)sin(10πf _(LO2) t)+(1/7)sin(14πf _(LO2) t)+ . . . ]  (40)

[0115] Therefore, the formula (31) is represented by $\begin{matrix}\begin{matrix}{{V_{IF}(t)} = \quad {u_{RF} \cdot U_{L01} \cdot {{sgn}\left( {S_{1}(t)} \right)} \cdot U_{L02} \cdot {{sgn}\left( {S_{2}^{\prime}(t)} \right)}}} \\{= \quad {{U_{RF} \cdot \cos}\quad \left( {2\quad \pi \quad f_{RF}t} \right){U_{L01} \cdot {{sgn}\left( {S_{1}(t)} \right)}}{U_{L02} \cdot {{sgn}\left( {S_{2}(t)} \right)}}}} \\{= \quad {\left( {16/\pi^{2}} \right){U_{RF} \cdot U_{L01} \cdot U_{L02} \cdot {{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}\left\lbrack {{\cos \left( {2\quad \pi \quad f_{L01}t} \right)} -} \right.}}}} \\{\quad {{\left( {1/3} \right){\cos \left( {6\quad \pi \quad f_{L01}t} \right)}} + {\left( {1/5} \right){\cos \left( {10\quad \pi \quad f_{L01}t} \right)}} -}} \\{\left. \quad {{\left( {1/7} \right){\cos \left( {14\quad \pi \quad f_{L01}t} \right)}} + \quad \cdots} \right\rbrack \times \left\lbrack {{- {\sin\left( {2\quad \pi \quad f_{L02}t} \right)}} +} \right.} \\{\quad {{\left( {1/3} \right){\sin \left( {6\quad \pi \quad f_{L02}t} \right)}} - {\left( {1/5} \right){\sin \left( {10\quad \pi \quad f_{L02}t} \right)}} +}} \\\left. \quad {{\left( {1/7} \right){\sin \left( {14\quad \pi \quad f_{L02}t} \right)}} + \quad \cdots} \right\rbrack \\{= \quad {{- \left( {16/\pi^{2}} \right)}{U_{RF} \cdot U_{L01} \cdot U_{L02} \cdot}}} \\{\quad \left\lbrack {{{\cos\left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {2\quad \pi \quad f_{L01}t} \right)}{\sin \left( {2\quad \pi \quad f_{L02}t} \right)}} +} \right.} \\{\quad {{\left( {1/9} \right){\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {6\quad \pi \quad f_{L01}t} \right)}{\sin \left( {6\quad \pi \quad f_{L02}t} \right)}} +}} \\{\quad {{\left( {1/25} \right){\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {10\pi \quad f_{L01}t} \right)}{\sin \left( {10\pi \quad f_{L02}t} \right)}} +}} \\{\quad {{\left( {1/49} \right){\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {14\quad \pi \quad f_{L01}t} \right)}{\sin \left( {14\quad \pi \quad f_{L02}t} \right)}} -}} \\{\quad {\left( {1/3} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {6\quad \pi \quad f_{L01}t} \right)}{\sin \left( {2\pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {2\quad \pi \quad f_{L01}t} \right)}{\sin \left( {6\quad \pi \quad f_{L02}t} \right)}} \right\} +} \\{\quad {\left( {1/5} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {10\quad \pi \quad f_{L01}t} \right)}{\sin \left( {2\quad \pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {2\quad \pi \quad f_{L01}t} \right)}{\sin \left( {10\quad \pi \quad f_{L02}t} \right)}} \right\} -} \\{\quad {\left( {1/7} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {14\quad \pi \quad f_{L01}t} \right)}{\sin \left( {2\quad \pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {2\quad \pi \quad f_{L01}t} \right)}{\sin \left( {14\pi \quad f_{L02}t} \right)}} \right\} -} \\{\quad {\left( {1/15} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {6\quad \pi \quad f_{L01}t} \right)}{\sin \left( {10\quad \pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {10\quad \pi \quad f_{L01}t} \right)}{\sin \left( {6\quad \pi \quad f_{L02}t} \right)}} \right\} -} \\{\quad {\left( {1/21} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {6\quad \pi \quad f_{L01}t} \right)}{\sin \left( {14\pi \quad f_{L02}t} \right)}} +} \right.}} \\{\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {14\pi \quad f_{L01}t} \right)}{\sin \left( {6\quad \pi \quad f_{L02}t} \right)}} \right\} -} \\{\quad {\left( {1/35} \right)\left\{ {{{\cos \left( {2\quad \pi \quad f_{RF}t} \right)}{\cos \left( {10\quad \pi \quad f_{L01}t} \right)}{\sin \left( {14\quad \pi \quad f_{L02}t} \right)}} +} \right.}} \\\left. {\left. \quad {\cos \quad \left( {2\quad \pi \quad f_{RF}t} \right){\cos \left( {14\quad \pi \quad f_{L01}t} \right)}{\sin \left( {10\quad \pi \quad f_{L02}t} \right)}} \right\} + \cdots} \right\rbrack\end{matrix} & (41)\end{matrix}$

[0116] From the formula (41), when the RF signal U_(RF) is switched bythe polarities of the local oscillator signals S₁(t) and S2′(t), basicwaves and harmonic waves of the three signals are obtained.

[0117] For example, four frequency components such assin{2π(f_(RF)−f_(LO1)−f_(LO2))t}, sin{2π(f_(RF)−f_(LO1)+f_(LO2))t},sin{2π(f_(RF)+f_(LO1)+f_(LO2))t} and sin{2π(f_(RF)+f^(LO1)−f_(LO2))t}are obtained from the triple product cos(2πf_(RF)t)cos(2πf_(LO1)t)sin(2πf_(LO2)t) of basic waves with reference to the formula of tripleproduct of trigonometric functions shown in the formula (17).

[0118] In this case, if the doubly-polarity switching mixer of FIG. 9constitutes a down-conversion system where a low-pass filter followsthis doubly-polarity switching mixer, only the low frequency componentsin{2π(f_(RF)−f_(LO1)−f_(LO2))t} is obtained. Note that a difference|f_(LO1)−f_(LO2)| is defined as an intermediate frequency f_(IF).

[0119] Also, four frequency components such assin[2π{f_(RF)−(2m+1)f_(LO1)−(2m′+1)f_(LO2)}t],sin[2π{f_(RF)−(2m+1)f_(LO1)+(2m′+1)f_(LO2)}t],sin[2π{f_(RF)+(2m+1)f_(LO1)+(2m′+1)f_(LO2)}t] andsin[2π{f_(RF)+(2m+1)f_(LO1)−(2m′+1)f_(LO2)}t] are obtained from tripleproducts cos(2πf_(RF)t) cos{2π(2m+1)f_(LO1)t}sin{2π(2m′+1)f_(LO2)t} (m,m′=1, 2, . . . ) of the RF signal V_(RF)(t) and the odd-higher harmonicwaves of the local oscillator signals S₁(t) and S₂(t) with reference tothe formula of triple product of trigonometric functions shown in theformula (17). In this case, these triple products decay with acoefficient of 1/{(2m+1) (2m′+1)}. Also, the frequencies of these tripleproducts are on the odd-higher order. For example, whenf_(LO1)=2nf_(LO2) (n=2, 3, . . . ) and f_(RF)=f_(LO1)+f_(LO2), theclosest frequency is 7f_(LO1)/4. Here, if f_(LO2)=f_(LO1)/2 and m=m′=1,3f_(LO1)/2=f_(RE).

[0120] In FIG. 10A, which illustrates a detailed circuit diagram of thedoubly-polarity switching mixer of FIG. 9, each of the switches 91, 92,93 and 94 are constructed by emitter-coupled pairs (current switches) ofbipolar transistors Q1, Q2; Q3, Q4; Q5, Q6; and Q7, Q8. In this case,one of the transistors Q1 and Q2, one of the transistors Q3 and Q4, oneof the transistors Q5 and Q6, and one of the transistors Q7 and Q8 areturned ON, while the other of the transistors Q1 and Q2, the other ofthe transistors Q3 and Q4, the other of the transistors Q5 and Q6, andthe other of the transistors Q7 and Q8 are turned OFF, so thatdifferential currents I_(RF) ⁺ and I_(RF) ⁻ generated from a lineardifferential circuit 101 by the REF signal V_(RF) are switched. Thus, anintermediate signal V_(IF) is obtained. Note that the doubly-polarityswitching mixer of FIG. 10A also serves as an analog circuit andaccordingly, serves as a three-input multiplier.

[0121] In FIG. 10B, which is a detailed circuit diagram of the lineardifferential circuit 101 of FIG. 10A, a current mirror circuit is formedby bipolar transistors Q9 and Q10 and another current mirror circuit isformed by bipolar transistors Q11 and Q12. Collectors of the bipolartransistors Q10 and Q11 are connected to emitters of bipolar transistorsQ13 and Q14, respectively, whose bases receive the RF signal V_(RF).Also, an emitter degeneration resistor R_(EE) is connected between theemitters of the bipolar transistors Q13 and Q14. Further, one currentsource I₀ is connected to each of the collectors of the bipolartransistors Q13 and Q14, and also, bases of bipolar transistors Q15 andQ16 serving as current sources to the current mirror circuits (Q9, Q10;Q11, Q12) are connected to the collectors of the bipolar transistors Q13and Q14, respectively.

[0122] In FIG. 10B, the following differential currents I_(RF) ⁺ andI_(RF) ⁻ are generated from the collectors of the bipolar transistors Q9and Q12.

I _(RF) ⁺ =I ₀ +V _(RF) /R _(EE)  (42)

I _(RF) ⁻ =I ₀ −V _(RF) /R _(EE)  (43)

[0123] That is, the linear characteristics of the linear differentialcircuit 101 are determined by the emitter degeneration resistor R_(EE).Note that a resistance manufactured by a semiconductor manufacturingprocess has excellent linear characteristics. Therefore, the lineardifferential circuit 101 has excellent linear characteristics.

[0124] In FIG. 11A, which illustrates another detailed circuit diagramof the doubly-polarity switching mixer of FIG. 9, each of the switches91, 92, 93 and 94 are constructed by source-coupled pairs (currentswitches) of MOS transistors M1, M2; M3, M4; M5, M6; and M7, M8. In thiscase, one of the transistors M1 and M2, one of the transistors M3 andM4, one of the transistors M5 and M6, and one of the transistors M7 andM8 are turned ON, while the other of the transistors M1 and M2, theother of the transistors M3 and M4, the other of the transistors M5 andM6, and the other of the transistors M7 and M8 are turned OFF, so thatdifferential currents I_(RF) ⁺ and I_(RF) ⁻ generated from a lineardifferential circuit 111 by the RF signal V_(RF) are switched. Thus, anintermediate signal V_(IF) is obtained. In the doubly-polarity switchingmixer of FIG. 11A, the transconductance does not change monotonously fora small signal change. Therefore, the doubly-polarity switching mixer ofFIG. 11A does not serve as an analog circuit and accordingly, does notserve as a three-input multiplier.

[0125] In FIG. 11B, which is a detailed circuit diagram of the lineardifferential circuit 111 of FIG. 11A, a current mirror circuit is formedby MOS transistors M9 and M10 and another current mirror circuit isformed by MOS transistors M11 and M12. Drains of the MOS transistors M10and M11 are connected to sources of MOS transistors M13 and M14,respectively, whose gates receives the RF signal V_(RF). Also, a sourcedegeneration resistor R_(EE) is connected between the sources of the MOStransistors M13 and M14. Further, one current source I₀ is connected toeach of the drains of the MOS transistors M13 and M14, and also, gatesof MOS transistors M15 and M16 serving as current sources to the currentmirror circuits (M9, M10; M11, M12) are connected to the drains of theMOS transistors M13 and M14, respectively.

[0126] Even in FIG. 11B, the differential currents I_(RF) ⁺ and I_(RF) ⁻represented by the formulae (42) and (43) are generated from the drainsof the MOS transistors M9 and M12.

[0127] That is, the linear characteristics of the linear differentialcircuit 111 are determined by the source degeneration resistor R_(EE).Note that a resistance manufactured by a semiconductor manufacturingprocess has excellent linear characteristics. Therefore, the lineardifferential circuit III has excellent linear characteristics.

[0128] Generally, in frequency mixers, the suppression of high-orderdistortion characteristics such as second-order and third-orderdistortion characteristics, i.e., second-order and third-order interceptpoint characteristics are important. In the doubly-polarity switchingmixers of FIGS. 10A and 11A, the high-order distortion characteristicscan be sufficiently suppressed by the linear differential circuit ofFIGS. 10B and 11B. However, in the doubly-polarity switching mixers ofFIGS. 10A and 11A, the power supply voltage V_(CC) or V_(DD) needs to behigher than 2V.

[0129] In FIG. 12, which illustrates a modification of thedoubly-polarity switching mixer of FIG. 10A, two triple tail cells C1and C2 are provided. In the triple tail cell C1, three emitter-coupledbipolar transistors Q1′, Q2′ and Q2′ are driven by one tail currentI_(RF) ⁺, while in the triple tail cell C2, three emitter-coupledbipolar transistors Q4′, Q5′ and Q6′ are driven by one tail currentI_(RF) ⁻. The tail currents I_(RF) ⁺ and I_(RF) ⁻ are generated by a V-Iconversion circuit 121 which can be easily constructed by bipolartransistors whose emitters are grounded via emitter resistors, where useis made of base voltage-to-collector current characteristics as V-Icharacteristics. In FIG. 12, the power supply voltage V_(CC) can belower than 1V.

[0130] In more detail, the transistors Q1′, Q2′, Q4′ and Q5′ areswitched by the local oscillator signal V_(LO1) or S₁(t), and thetransistors Q3′ and Q6′ are switched by the local oscillator signalV_(LO2) or S₂(t). In this case, when the transistor Q3′ is turned ON bythe local oscillator signal V_(LO2), the tail current I_(RF) ⁺ needs tobe supplied from the power supply voltage V_(CC) regardless of whetherthe transistors Q1′ and Q2′ are turned ON or OFF. On the other hand,when the transistor Q6′ is turned ON by the local oscillator signalV_(LO2), the tail current I_(RF) ⁻ needs to be supplied from the powersupply voltage V_(CC) regardless of whether the transistors Q4′ and Q5′are turned ON or OFF. For this purpose, the transistors Q3′ and Q6′ areincreased in size or the amplitude of the local oscillator signalV_(LO2) is larger than that of the local oscillator signal V_(LO1).

[0131] Thus, the doubly-polarity switching mixer of FIG. 12 isequivalent to the doubly-polarity switching mixer of FIGS. 10A and 10Band has an advantage in that the power supply voltage V_(CC) is low.

[0132] In FIG. 13, which illustrates a modification of thedoubly-polarity switching mixer of FIG 11A, two triple tail cells C1 andC2 are provided. In the triple tail cell C1, three source-coupled MOStransistors M1′, M2′ and M2′ are driven by one tail current I_(RF) ⁺,while, in the triple tail cell C2, three emitter-coupled MOS transistorsM4′, M5′ and M6′ are driven by one tail current I_(RF) ⁻. The tailcurrents I_(RF) ⁺ and I_(RF) ⁻ are generated by a V-I conversion circuit131 which can be easily constructed by MOS transistors whose sources aregrounded via source resistors, where use is made of gatevoltage-to-drain current characteristics as V-I characteristics. In FIG.13, the power supply voltage V_(DD) can be lower than 1V.

[0133] In more detail, the transistors M1′, M2′, M4′ and M5′ areswitched by the local oscillator signal V_(LO1) or S₁(t), and thetransistors M3′ and M6′ are switched by the local oscillator signalV_(LO2) or S₂(t). In this case, when the transistor M3′ is turned ON bythe local oscillator signal V_(LO2), the tail current I_(RF) ⁺ needs tobe supplied from the power supply voltage V_(DD) regardless of whetherthe transistors M1′ and M2′ are turned ON or OFF. On the other hand,when the transistor M6′ is turned ON by the local oscillator signalV_(LO2), the tail current I_(RF) ⁻ needs to be supplied from the powersupply voltage V_(DD) regardless of whether the transistors M4′ and M5′are turned ON or OFF. For this purpose, the transistors M3′ and M6′ areincreased in size or the amplitude of the local oscillator signalV_(LO2) is larger than that of the local oscillator signal V_(LO1).

[0134] Thus, the doubly-polarity switching mixer of FIG. 13 isequivalent to the doubly-polarity switching mixer of FIGS. 11A and 11Band has an advantage in that the power supply voltage V_(DD) is low.

[0135] In FIG. 14A, which illustrates a modification of thedoubly-polarity switching mixer of FIG. 12, two triple tail cells C3 andC4 are added, and a dual linear differential circuit 141 is providedinstead of the V-I conversion circuit 121 of FIG. 12. The dual lineardifferential circuit 141 which is similar to the linear differentialcircuit 101 of FIG. 10B is illustrated in detail in FIG. 14B. In thetriple tail cell C3, three emitter-coupled bipolar transistors Q7′, Q8′and Q9′ are driven by one tail current V_(RF) ⁺, and in the triple tailcell C4, three emitter-coupled bipolar transistors Q10′, Q11′ and Q12′are driven by one tail current I_(RF) ⁻. In this case, the intermediatesignal V_(IF) is obtained by a difference between a sum current flowingthrough the transistors Q1′, Q5′, Q8′ and Q10′ and a sum current flowingthrough the transistors Q2′, Q4′, Q7′ and Q11′.

[0136] In FIG. 15A, which illustrates a modification of thedoubly-polarity switching mixer of FIG. 13, two triple tail cells C3 andC4 are added, and a dual linear differential circuit 151 is providedinstead of the V-I conversion circuit 131 of FIG. 13. The dual lineardifferential circuit 151 which is similar to the linear differentialcircuit 101 of FIG. 11B is illustrated in detail in FIG. 15B. In thetriple tail cell C3, three emitter-coupled bipolar transistors M7′, M8′and M9′ are driven by one tail current V_(RF) ⁺, and in the triple tailcell C4, three emitter-coupled bipolar transistors M10′, M11′ and M12′are driven by one tail current I_(RF) ⁺. In this case, the intermediatesignal V_(IF)is obtained by a difference between a sum current flowingthrough the transistors M1′, M5′, MS′ and M10′ and a sum current flowingthrough the transistors M2′, M4′, M7′ and M11′.

[0137] In FIG. 16, which illustrates a third embodiment of thequadrature mixer circuit applied to a direct conversion type wirelessreceiver according to the present invention, a 1/2n(n=1, 2, . . . )frequency divider 161 is provided instead of the ½-frequency divider 71and 72 of FIG. 7. In the ¼-frequency divider 161 where n=1 isconstructed by a Johnson counter, the quadrature mixer circuit of FIG.16 is the same as that of FIG. 7.

[0138] In FIG. 16,

f _(RF)=(2n+1)f _(LO)/(2n)

[0139] Also, since f_(LO2)=f_(LO)/(2n) the frequency fuzz of the outputsignals of the ½n-frequency divider 161 is represented by

f _(RF)=(2n+1)f _(LO2)  (44)

[0140] As apparent from the formulae (28) and (38), since the outputsignals of the ½n-frequency divider 161 are rectangular, odd-higherorder harmonic frequencies (2j−1) f_(LO2) (j=2, 3, . . . ) are includedtherein. From the formula (44), some of such harmonic frequencies alwayscoincide with the frequency f_(RF) of the RF signal V_(RF), which wouldincrease the DC offset and the reception trouble as in the prior art.

[0141] In FIG. 17, which illustrates a fourth embodiment of thequadrature mixer circuit applied to a direct conversion type wirelessreceiver according to the present invention, a 1/m-frequency divider 171is connected to the voltage controlled oscillator 33″ to generate afirst local oscillator signal u_(LO1), and a 1/m′-frequency divider 172is connected to the voltage controlled oscillator 33″ to generate asecond local oscillator signal u_(LO2). In this case,

f _(LO1) =f _(LO) /m

f _(LO2) =f _(LO) /m′

[0142] then,

f _(RF) =f _(LO1) +f _(LO2) =f _(LO) /m+f _(LO) /m′=(m+m′)/(m m′)·f_(LO)  (45)

[0143] The frequency component of the first local oscillator signalu_(LO1) includes (2i−1) f_(LO)/m (i=1, 2, . . . ) and the frequencycomponent of the second local oscillator signal u_(LO2) includes (2j−1)f_(LO)/m′(j=1, 2, . . . ). These frequencies should not coincide withthe frequency f_(RF) of the RF signal. That is,

(m+m′)/(m m′)≠1(2i−1)/m≠(m+m′)/(m m′)(2j−1)/m′≠(m+m′)/(m m′)

[0144] In other words,

1/m+1/m′≠1  (46)

i≠m/(2m′)+1  (47)

j≠m′/(2m)+1  (48)

[0145] From the formulae (47) and (48), i. j=2, 3, . . .

[0146] The values of m and m′ satisfying the formulae (46), (47) and(48) are shown in FIG. 18. In this case, any even-ordered high orderwaves of the local oscillator signals do not coincide with the frequencyf_(RF) of the RF signal. Also, in any cases, f_(LO>f) _(RF). Forexample, the values m and m′ satisfying that f_(RF)<f_(LO)<2f_(RF) areas follows:

f _(LO)=6f _(RF)/5 at (m, m′)=(3, 2)

f _(LO)=¹⁰ _(f) _(RF)/7 at (m, m′)=(5, 2)

f _(LO)=14f _(RF)/9 at (m, m′)=(7, 2)

f _(LO)=18f _(RF)/11 at (m, m′)=(9, 2)

f _(LO)=5f _(RF)/3 at (m, m′)=(10, 2)

f _(LO)=22f _(RF)/13 at (m, m′)=(11, 2)

f _(LO)=26f _(RF)/15 at (m, m′)=(13, 2)

f _(LO)=7f _(RF)/4 at (m, m′)=(14, 2)

f _(LO)=52f _(RF)/17 at (m, m′)=(15, 2)

f _(LO)=12f _(RF)/7 at (m, m′)=(3, 4)

f _(LO)=3f _(RF)/2 at (m, m′)=(2, 6)

[0147] Also, the values of m and m′ satisfying thatf_(RF)<f_(LO)<1.5f_(RF) are as follows:

f _(LO)=6f _(RF)/5 at (m, m′)=(3, 2)

f _(LO)=10f _(RF)/7 at (m, m′)=(5, 2)

[0148] In FIG. 17, if the 1/m′-frequency divider 172 includes a Johnsoncounter (½-frequency divider),

m′=2n(n=1, 2, . . . )

[0149] Then,

f _(RF)=(m+2n)(2mn)·f _(LO)  (50)

[0150] Even in this case, the frequency component of the first localoscillator signal u_(LO1) includes (2i−1) f_(LO)/m (i=1, 2, . . . ) andthe frequency component of the second local oscillator signal u_(LO2)includes (2j−1) f_(LO)/(2n)(j=1, 2, . . . ). These frequencies shouldnot coincide with the frequency f_(RF) of the RF signal. That is,

(m+2n)/(2m n)≠1

(2i−1)/m≠(m+2n)/(2m n)

(2j−1)/2n ≠( m+2n)/(2m n)

[0151] In other words,

1/m+1/2n≠1  (51)

i≠m/(4n)+1  (52)

j≠n/m+1  (53)

[0152] From the formulae (52) and (53), i. j=2, 3, . . .

[0153] The values of m and 2n satisfying the formulae (51), (52) and(53) are shown in FIG. 19. In this case, any even-ordered high orderwaves of the local oscillator signals do not coincide with the frequencyf_(RF) of the RF signal. Also, in any cases, f_(LO)>f_(RF). For example,the values m and n satisfying that f_(RF)<f_(LO)<2f_(RF) are as follows:

f _(LO)=6f _(RF)/5 at (m, n)=(3, 1)

f _(LO)=10f _(RF)/7 at (m, n)=(5, 1)

f _(LO)=14f _(RF)/9 at (m, n)=(7, 1)

f _(LO)=18f _(RP)/11 at (m, n)=(9, 1)

f _(LO)=5f _(RF)/3 at (m, n)=(10, 1)

f _(LO)=22f _(RF)/13 at (m, n)=(11, 1)

f _(LO)=26f _(RF)/15 at (m, n)=(13, 1)

f _(LO)=7f _(RF)/4 at (m, n)=(14, 1)

f _(LO)=52f _(RF)/17 at (m, n)=(15, 1)

f _(LO)=12f _(RF)/7 at (m, n)=(3, 2)

f _(LO)=3f _(RF)/2 at (m, n)=(2, 3)

[0154] Also, the values of m and m′ satisfying thatf_(RF)<f_(LO)<1.5f_(RF) are as follows:

f _(LO)=6f _(RF)/5 at (m, n)=(3, 1)

_(fLO)=10f _(RF)/7 at (m, n)=(5, 1)

[0155] According to the inventor's calculation the noise factor (NF) ofthe quadrature mixer circuit according to the present invention wasabout 7 dB while the NF of the first prior art quadrature mixer circuitas illustrated in FIG. 1 was 3.01 dB. Such deterioration of the NF canbe compensated for by increasing the power gain of the AGC amplifiers 6and 7 in a direct conversion type wireless receiver, and accordingly,there is no actual problem.

[0156] Also, the present invention can be applied to a low IF typewireless receiver.

[0157] As explained hereinabove, according to the present invention, theDC offset can be decreased, the reception trouble can be suppressed, andthe number of components can be decreased.

1. A quadrature mixer circuit for receiving a radio frequency signal togenerate first and second quadrature output signals, comprising: a firstthree-input mixer for receiving said radio frequency signal, a firstlocal signal having a first frequency and a second local signal having asecond frequency to generate said first quadrature output signal; and asecond three-input mixer for receiving said radio frequency signal, saidfirst local signal and said second local signal to generate said secondquadrature output signal, said second local signal received by saidfirst three-input mixer and said second local signal received by saidsecond three-input mixer being out of phase by π/2 from each other. 2.The quadrature mixer circuit as set forth in claim 1, furthercomprising: a first oscillator for generating said first local signal; asecond oscillator for generating said second local signal; and a π/2phase shifter, connected to said second oscillator, for shifting a phaseof said second local signal by π/2, an output signal of said secondoscillator being supplied to one of said first and second three-inputmixers, an output signal of said π/2 phase shifter being supplied to theother of said first and second three-input mixers.
 3. The quadraturemixer circuit as set forth in claim 1, further comprising: an oscillatorfor generating said first local signal; and a frequency divider,connected to said oscillator, for generating said second local signal.4. The quadrature mixer circuit as set forth in claim 3, wherein saidfrequency divider comprises a ½-frequency divider for generating saidsecond local signal and a phase-shifted signal of said second localsignal by a phase of π/2.
 5. The quadrature mixer circuit as set forthin claim 1, further comprising: an oscillator; a 1/m-frequency divider,connected to said oscillator, for generating said first local signal; a1/m′-frequency divider, connected to said oscillator, for generatingsaid second local oscillator.
 6. The quadrature mixer circuit as setforth in claim 5, wherein the values of m and m′ satisfy:1/m+1/m′≠1,i≠m/(2m′)+1, andj≠m′/(2m)+1 where i,j=2,3, . . .
 7. Thequadrature mixer circuit as set forth in claim 1, further comprising: anoscillator; a 1/m-frequency divider, connected to said oscillator, forgenerating said first local signal; a 1/(2n)-frequency divider,connected to said oscillator, for generating said second localoscillator.
 8. The quadrature mixer circuit as set forth in claim 7,wherein the values of m and n satisfy: 1/m+1/(2n)≠1,i≠m/(4n)+1,andj≠n/m+1′where i,j=2, 3, . . .
 9. The quadrature mixer circuit as setforth in claim 1, wherein each of said three-input mixers comprises athree-input multiplier.
 10. The quadrature mixer circuit as set forth inclaim 1, wherein each of said three-input mixers comprises adoubly-polarity switching mixer.
 11. The quadrature mixer circuit as setforth in claim 10, wherein said first and second local signals arerectangular.
 12. The quadrature mixer circuit as set forth in claim 10,wherein said doubly-polarity switching mixer comprises non-linearelements showing three-order transfer characteristics for said radiofrequency signal, said first local signal and said second local signal.13. The quadrature mixer circuit as set forth in claim 12, wherein saidnon-linear elements comprise bipolar transistors.
 14. The quadraturemixer circuit as set forth in claim 12, wherein said non-linear elementscomprise MOS transistors.
 15. The quadrature mixer circuit as set forthin claim 10, wherein said doubly-polarity switching mixer comprises: alinear differential circuit for generating first and second differentialcurrents in response to said radio frequency signal; first and secondbipolar transistors having a common emitter for receiving said firstdifferential current, said first local signal being supplied betweenbases of said first and second bipolar transistors; third and fourthbipolar transistors having a common emitter for receiving said seconddifferential current, said first local signal being supplied betweenbases of said third and fourth bipolar transistors; fifth and sixthbipolar transistors having a common emitter connected to a commoncollector of said first and third bipolar transistors, said second localsignal being supplied between bases of said fifth and sixth bipolartransistors; seventh and eighth bipolar transistors having a commonemitter connected to a common collector of said second and fourthbipolar transistors, said second local signal being supplied betweenbases of said seventh and eighth bipolar transistors; a respective oneof said first and second quadrature output signals being a differencebetween a sum current flowing through said sixth and eighth bipolartransistors and a sum current flowing through said fifth and seventhbipolar transistors.
 16. The quadrature mixer circuit as set forth inclaim 10, wherein said doubly-polarity switching mixer comprises: alinear differential circuit for generating first and second differentialcurrents in response to said radio frequency signal; first and secondMOS transistors having a common source for receiving said firstdifferential current, said first local signal being supplied betweengates of said first and second MOS transistors; third and fourth MOStransistors having a common source for receiving said seconddifferential current, said first local signal being supplied betweengates of said third and fourth MOS transistors; fifth and sixth MOStransistors having a common source connected to a common drain of saidfirst and third MOS transistors, said second local signal being suppliedbetween gates of said fifth and sixth MOS transistors; seventh andeighth MOS transistors having a common source connected to a commondrain of said second and fourth MOS transistors, said second localsignal being supplied between gates of said seventh and eighth MOStransistors; a respective one of said first and second quadrature outputsignals being a difference between a sum current flowing through saidsixth and eighth MOS transistors and a sum current flowing through saidfifth and seventh MOS transistors.
 17. The quadrature mixer circuit asset forth in claim 10, wherein said doubly-polarity switching mixercomprises; a voltage-to-current conversion circuit for converting saidradio frequency signal into first and second differential currents;first, second and third bipolar transistors having a common emitter forreceiving said first differential current, said first local signal beingsupplied between bases of said first and second bipolar transistors, acollector of said third bipolar transistor being connected to a powersupply terminal; and fourth, fifth and sixth bipolar transistors havinga common emitter for receiving said second differential current, saidfirst local signal being supplied between bases of said fourth and fifthbipolar transistors, a collector of said sixth bipolar transistor beingconnected to said power supply terminal, said second local signal beingsupplied between bases of said third and sixth bipolar transistors, arespective one of said first quadrature output signals being adifference between a sum current flowing through said first and fifthbipolar transistors and a sum current flowing through said second andfourth bipolar transistors.
 18. The quadrature mixer circuit as setforth in claim 10, wherein said doubly-polarity switching mixercomprises: a voltage-to-current conversion circuit for converting saidradio frequency signal into first and second differential currents;first, second and third MOS transistors having a common source forreceiving said first differential current, said first local signal beingsupplied between gates of said first and second MOS transistors, a drainof said third MOS transistor being connected to a power supply terminal;and fourth, fifth and sixth MOS transistors having a common source forreceiving said second differential current, said first local signalbeing supplied between gates of said fourth and fifth MOS transistors, adrain of said sixth MOS transistor being connected to said power supplyterminal, said second local signal being supplied between gates of saidthird and sixth MOS transistors, a respective one of said firstquadrature output signals being a difference between a sum currentflowing through said first and fifth MOS transistors and a sum currentflowing through said second and fourth MOS transistors.
 19. Thequadrature mixer circuit as set forth in claim 10, wherein saiddoubly-polarity switching mixer comprises: a dual linear differentialcircuit for generating first, second, third and fourth differentialcurrents, said first and second differential currents being similar toeach other, said third and fourth differential currents being similar toeach other; first, second and third bipolar transistors having a commonemitter for receiving said first differential current, said first localsignal being supplied between bases of said first and second bipolartransistors, a collector of said third bipolar transistor beingconnected to a power supply terminal; fourth, fifth and sixth bipolartransistors having a common emitter for receiving said seconddifferential current, said first local signal being supplied betweenbases of said fourth and fifth bipolar transistors, a collector of saidsixth bipolar transistor being connected to said power supply terminal;seventh, eighth and ninth bipolar transistors having a common emitterfor receiving said third differential current, said first local signalbeing supplied between bases of said seventh and eighth bipolartransistors, a collector of said ninth bipolar transistor beingconnected to said power supply terminal; and tenth, eleventh and twelfthbipolar transistors having a common emitter for receiving said fourthdifferential current, said first local signal being supplied betweenbases of said tenth and eleventh bipolar transistors, a collector ofsaid twelfth bipolar transistor being connected to said power supplyterminal; said second local signal being supplied between bases of saidthird and sixth bipolar transistors, said second local signal beingsupplied between bases of said ninth and twelfth bipolar transistors, arespective one of said first quadrature output signals being adifference between a sum current flowing through said first, fifth,eighth and tenth bipolar transistors and a sum current flowing throughsaid second, fourth, seventh and eleventh bipolar transistors.
 20. Thequadrature mixer circuit as set forth in claim 10, wherein saiddoubly-polarity switching mixer comprises: a dual linear differentialcircuit for generating first, second, third and fourth differentialcurrents, said first and second differential currents being similar toeach other, said third and fourth differential currents being similar toeach other; first, second and third MOS transistors having a commonsource for receiving said first differential current, said first localsignal being supplied between gates of said first and second MOStransistors, a drain of said third MOS transistor being connected to apower supply terminal; fourth, fifth and sixth MOS transistors having acommon source for receiving said second differential current, said firstlocal signal being supplied between gates of said fourth and fifth MOStransistors, a drain of said sixth MOS transistor being connected tosaid power supply terminal; seventh, eighth and ninth MOS transistorshaving a common source for receiving said third differential current,said first local signal being supplied between gates of said seventh andeighth MOS transistors, a drain of said ninth MOS transistor beingconnected to said power supply terminal; and tenth, eleventh and twelfthMOS transistors having a common source for receiving said fourthdifferential current, said first local signal being supplied betweengates of said tenth and eleventh MOS transistors, a drain of saidtwelfth MOS transistor being connected to said power supply terminal;said second local signal being supplied between gates of said third andsixth MOS transistors, said second local signal being supplied betweengates of said ninth and twelfth MOS transistors, a respective one ofsaid first quadrature output signals being a difference between a sumcurrent flowing through said first, fifth, eighth and tenth MOStransistors and a sum current flowing through said second, fourth,seventh and eleventh MOS transistors.
 21. The quadrature mixer circuitas set forth in claim 1, applied to a direct conversion type wirelessreceiver.
 22. The quadrature mixer circuit as set forth in claim 1,applied to a low intermediate frequency type wireless receiver.